Nyquist folded bandpass sampling receivers with narrow band filters for UWB pulses and related methods

ABSTRACT

Nyquist folded bandpass sampling receivers are disclosed that utilize narrow band filters in parallel with wideband filters to enhance reception of ultra wideband (UWB) pulses. The addition of the narrow band filter facilitates the reception of ultra wideband signal pulses and, therefore, extends the Nyquist folding bandpass sampling receiver to allow improved processing of ultra wideband (UWB) pulses. RF sampling circuitry utilizing a modulated sampling clock signal can then better capture UWB pulse signals.

RELATED APPLICATIONS

This application is a continuation-in-part application of the followingtwo co-pending applications: U.S. patent application Ser. No. 11/247,338entitled “RECONFIGURABLE DIRECT RF BANDPASS SAMPLING RECEIVER ANDRELATED METHODS,” which was filed on Oct. 11, 2005, and U.S. patentapplication Ser. No. 11/247,314 entitled “SWEPT BANDPASS FILTERFREQUENCY MODULATED CONTINUOUS WAVE (FMCW) RECEIVER AND RELATEDMETHODS,” which was filed on Oct. 11, 2005, the entire text and allcontents for each of which is hereby expressly incorporated by referencein its entirety.

This application is also related in part to the subject matter describedin the following concurrently filed patent applications: U.S. patentapplication Ser. No. 11/545,310, entitled “DIRECT BANDPASS SAMPLINGRECEIVERS WITH ANALOG INTERPOLATION FILTERS AND RELATED METHODS,” andU.S. patent application Ser. No. 11/545,642, entitled “NYQUIST FOLDEDBANDPASS SAMPLING RECEIVERS AND RELATED METHODS,” the entire text andall contents for each of which is hereby expressly incorporated byreference in its entirety.

TECHNICAL FIELD OF THE INVENTION

This invention relates to receiver architectures for the efficientsampling of radio frequency (RF) signals and, more particularly, toreceiver architectures for the efficient sampling of signals over a widefrequency range of interest.

BACKGROUND

A wide variety of signals and related protocols exist for the use ofradio frequency (RF) signals in communication systems and other devices,such as radar systems. In some applications, it is desirable todetermine or confirm the existence of RF signals and accurately measuretheir parameters. RF signals of interest, however, can occur across awide range of center frequencies with various bandwidths and can haverelatively small signals compared to background noise. As such, it isdesirable for an RF receiver to be designed to acquire and allow thedetection and measurement of signals across a wide frequency range withvarious bandwidths while contributing little distortion, spurs orinterference from its own circuitry. For a electronic intelligenceapplication, for example, the desired signals to be acquired anddetected can fall within a frequency range from less than 2 GHz togreater then 20 GHz. To provide reasonable sensitivity against a varietyof signal types and bandwidths while maximizing search coverage, typicalinstantaneous search bandwidths may range from 100 MHz or less to 1 GHzor greater.

Many receiver architectures currently exist for receiving and detectingRF signals. These architectures include heterodyne receivers, homodynereceivers (also called zero-IF and direct conversion receivers forintermediate frequency (IF) applications), low-IF receivers, doubleconversion wideband IF receivers, wideband digital receivers, 6-portreceivers (a special case of homodyne receivers), 3-phase variations ofhomodyne receivers, charge-domain direct RF mixer-sampler receivers,compressive receivers, noise-shaping sigma-delta receivers,non-reconfigurable direct RF optical down-sampling receivers, bandpasssampling variations of heterodyne receivers, and optical tunedchannelized filters for fiberoptic WDM (wavelength division multiplexed)receivers. In addition, multi-signal bandpass sampling receiverscombining the outputs from multiple bandpass filters without tuning havebeen proposed. In addition, noise-shaping sigma delta converters thatuse a bank of bandpass filters to implement a tuning function with amodulation sampling clock meeting the Nyquist criteria for the totalfrequency range of interest have been designed. In addition, direct RFreceivers based on the use of analog high-speed pre-samplers have beenbuilt, although not in any reconfigurable architecture. Still further,combination architectures have been utilized such as a combination ofswitched homodyne receiver and low-IF receiver architectures.

For wideband applications, sampling at the Nyquist rate of at leasttwice the bandwidth can be very difficult because of device limitations,power consumption, size, weight, and cost. In order to avoid thesedifficulties, sub-Nyquist sampling schemes have been proposed includingvarious non-uniform sampling techniques for harmonic retrieval and somerecent methods in compressive sensing (also referred to as compressivesampling). Non-uniform sampling techniques proposed to date have,however, been limited in the types of signals that can be processed(generally extremely narrow-band signals), number of simultaneoussignals (one or two typically), and total decimation ratio (typically1/5 to 1/10 Nyquist at best). Compressive sensing techniques suffer fromnumerous challenges, including device implementation, computationalcomplexity, and signal reconstruction.

Each of these prior architectures suffer certain disadvantages and,therefore, have not been entirely effective in receiving and detectingRF signals, particularly in applications requiring reconfigurability forvariable signal environment; the ability to reconstruct the signal;reasonable sensitivity; low size, weight, cost, and power; largefrequency range of interest that may span many GHz; includingapplications such as radar warning receivers, electronic supportreceivers, electronic support measures receivers, electronicintelligence, communications intelligence, and ultra wideband radarreceiver applications.

SUMMARY OF THE INVENTION

Nyquist folded bandpass sampling receivers are disclosed that utilizenarrow band filters in parallel with wideband filters to enhancereception of ultra wideband (UWB) pulses. The addition of the narrowband filter facilitates the reception of ultra wideband signal pulsesand, therefore, extends the Nyquist folding bandpass sampling receiverto allow improved processing of ultra wideband (UWB) pulses. RF samplingcircuitry utilizing a modulated sampling clock signal can then bettercapture UWB pulse signals. As described in more detail below, a varietysystems and methods can be utilized as part of the present invention.

In one embodiment, receive path circuitry is disclosed for a bandpasssampling receiver having folded Nyquist zones and ultra wideband pulsecapabilities including wideband filter circuitry receiving an RF inputsignal that has a center frequency within a frequency range of interestand has a bandwidth less than or equal to the frequency range ofinterest and wide enough to cover multiple Nyquist zones associated witha modulated sampling clock, narrow band filter circuitry receiving theRF input signal and having a center frequency within a frequency rangeof interest and having a bandwidth less than the Nyquist bandwidth ofthe modulated sampling clock, and sampling circuitry configured toreceive a filtered signal from the wideband filter circuitry, to receivea filtered signal from the narrow band filter circuitry, and to receivethe modulated sampling clock signal as an input where the modulatedsampling clock signal is configured to provide non-uniform sampling forsignals within the multiple Nyquist zones. In addition, the modulatedsampling clock signal can be a frequency modulated clock signal. Inaddition, the frequency modulated clock signal may include a linearsawtooth modulation, a sinusoidal modulation, a triangle modulation, afrequency shift key modulation, a frequency agile modulation, acommunications frequency modulation, or a combination thereof.

In a further embodiment for the receive path circuitry, the samplingcircuitry can include non-quantizing sampling circuitry, an analoginterpolation filter coupled to receive the output of the non-quantizingsampling circuitry where the analog interpolation filter has a centerfrequency within a Nyquist zone of operation for the non-quantizingsampling circuitry, and analog to digital converter (ADC) circuitryconfigured to receive a quantization sampling clock signal and toquantize an analog signal received from the analog interpolation filter.In addition, the analog interpolation filter can include a tunable orswitchable interpolation filter.

In a further embodiment, the modulated sampling clock can include afrequency modulated sampling clock having a tunable frequency or anadjustable modulation or both. Still further, the wideband filtercircuitry can include a wideband filter having bandwidth of 10 GHz ormore; the modulated sampling clock can have a sampling rate of 1 GHz ormore; and the narrow band filter circuitry can have a bandwidth lessthan 1 GHz. In addition, the wideband filter circuitry can include awideband filter having bandwidth of 20 GHz or more; the modulatedsampling clock can have a sampling rate of 2 GHz or more; and the narrowband filter circuitry can have a bandwidth less than 2 GHz. Also, thewideband filter circuitry can be a tunable bandpass filter having atunable center frequency dependent upon a filter control signal wherethe center frequency is tunable across the frequency range of interestand where the tunable bandpass filter has a bandwidth less than thefrequency range of interest. The narrow band filter circuitry caninclude one or more tunable narrow band filters each having a tunablecenter frequency dependent upon a filter control signal and each beingtunable across the frequency range of interest. And the receive pathcircuitry can further include digital signal processing circuitrycoupled to receive an output from the sampling circuitry and configuredto identify ultra wideband pulses based upon signals passed through thenarrow band filter circuitry.

In another embodiment, a method is disclosed for bandpass sampling ofsignals using folded Nyquist zones including utilizing wideband filtercircuitry to filter an RF input signal within a frequency range ofinterest where the bandpass filter has a bandwidth less than or equal tothe frequency range of interest and wide enough to cover multipleNyquist zones associated with a modulated sampling clock, also filteringthe RF input signal with narrow band filter circuitry having a centerfrequency within a frequency range of interest and having a bandwidthless than the Nyquist bandwidth of the modulated sampling clock,combining an output signal from the wideband filter circuitry with anoutput signal from the narrow band filter circuitry to generate acombined filtered signal, generating a modulated sampling clockconfigured to provide non-uniform sampling for signals within themultiple Nyquist zones, and bandpass sampling the combined filteredsignal from the wideband filter circuitry covering multiple Nyquistzones using the modulated sampling clock. In addition, the generatingstep can include generating a chirp frequency modulated clock signal forthe modulated sampling clock signal.

In another embodiment for the method, the bandpass sampling step caninclude bandpass sampling the combined filtered signal withoutquantizing the signal, filtering the sampled signal with an analoginterpolation filter having a center frequency within a Nyquist zone,and quantizing a signal received from the analog interpolation filter.In addition, the generating step can include generating a frequencymodulated clock signal for the modulated sampling clock signal, and themethod can include using a tunable or switchable interpolation filter asthe analog interpolation filter. Still further, the utilizing step caninclude utilizing wideband filter circuitry having a bandwidth of 10 GHzor more, the generating step can include generating a modulated samplingclock signal having a sampling rate of 1 GHz or more, and the filteringstep can include filtering with narrow band filter circuitry having abandwidth less than 1 GHz. In addition, the utilizing step can includeutilizing wideband filter circuitry having a bandwidth of 20 GHz ormore, the generating step can include generating a modulated samplingclock signal having a sampling rate of 2 GHz or more, and the filteringstep can include filtering with narrow band filter circuitry having abandwidth less than 2 GHz.

In a further embodiment, the utilizing step can include utilizing one ormore tunable bandpass filters and further comprising tuning the centerfrequency of the tunable bandpass filter within the frequency range ofinterest. In addition, the filtering step can include also filtering theRF input signal with one or more tunable narrow band filters and furthercomprising tuning the center frequency of the one or more tunable narrowband filters within the frequency range of interest. Still further, themethod can include processing sampled signals to identify ultra widebandpulses based upon signals passed through the narrow band filtercircuitry.

DESCRIPTION OF THE DRAWINGS

It is noted that the appended drawings illustrate only exemplaryembodiments of the invention and are, therefore, not to be consideredlimiting of its scope, for the invention may admit to other equallyeffective embodiments.

FIG. 1 is a block diagram of an embodiment for a reconfigurable directradio-frequency (RF) sampling receiver.

FIG. 2A is a block diagram for an example embodiment of a reconfigurabledirect RF bandpass sampling receiver with an analog interpolationfilter.

FIG. 2B is a Nyquist zone diagram related to the operation of the analoginterpolation filter.

FIG. 3 is a block diagram for a Nyquist folded bandpass samplingreceiver.

FIG. 4 is a block diagram for a Nyquist folded bandpass samplingreceiver having a chirped sample clock signal and a switchableinterpolation filter.

FIG. 5A is signal diagram for a modulated RF sampling clock using chirpor ramp modulation.

FIG. 5B is signal diagram for instantaneous frequency of two receivedsignals using the modulated RF sampling clock of FIG. 5A.

FIG. 6A is a Nyquist zone diagram for three signals received when themodulated RF sampling clock signal is at a first frequency.

FIG. 6B is a Nyquist zone diagram for three signals received when themodulated RF sampling clock signal is at a second frequency.

FIG. 7 is a block diagram for a Nyquist folded bandpass samplingreceiver that does not use an analog interpolation filter to de-couplethe RF sampling clock and the quantization clock.

FIG. 8 is a block diagram for a Nyquist folded bandpass samplingreceiver having a narrow band filter to improve reception of ultrawideband pulsed signals.

FIG. 9A is a block diagram for a Nyquist folded bandpass samplingreceiver having a modulated and possibly tunable sampling clock.

FIG. 9B is a block diagram for a Nyquist folded bandpass samplingreceiver with a tunable and modulated sampling clock and a front-endfilter that provides some anti-aliasing.

FIG. 10A is a block diagram for a Nyquist folded bandpass samplingreceiver utilizing an injected pilot tone to facilitate clock recovery.

FIG. 10B is a block diagram for example processing that could be used torecover the clock signal and analyze the signal data.

DETAILED DESCRIPTION OF THE INVENTION

Nyquist folded bandpass sampling receivers are disclosed that utilizewideband filters and modulated sampling clocks to identify receivedsignals. By using one or more clock modulations to induce frequencymodulations that are Nyquist zone dependent, multiple Nyquist zones canbe aliased together while still allowing for signals from differentNyquist zones to be separated and identified. Further, the addition ofthe narrow band filter facilitates the reception of ultra widebandsignal pulses and, therefore, extends the Nyquist folding bandpasssampling receiver to allow improved processing of ultra wideband (UWB)pulses. As described in more detail below, a variety systems and methodscan be utilized as part of the present invention.

Initially, a reconfigurable direct radio frequency (RF) bandpasssampling receiver is discussed with respect to FIG. 1. Thisreconfigurable direct radio frequency (RF) bandpass sampling receiver isalso described in U.S. patent application Ser. No. 11/247,338 entitled“RECONFIGURABLE DIRECT RF BANDPASS SAMPLING RECEIVER AND RELATEDMETHODS,” which was filed on Oct. 11, 2005, which is hereby incorporatedby reference in its entirety. Associated receiver architectures thatadditional take advantage of an analog interpolation filter are thendiscussed with respect to FIGS. 2A and 2B. This addition of an analoginterpolation filter and further embodiments are discussed in moredetail in the concurrently filed U.S. patent application Ser. No.11/545,310, entitled “DIRECT BANDPASS SAMPLING RECEIVERS WITH ANALOGINTERPOLATION FILTERS AND RELATED METHODS,” which is hereby incorporatedby reference in its entirety. Nyquist folded bandpass sampling receiversare then discussed with respect to FIGS, 3, 4, 5A, 5B, 6A, 6B and 7. Anembodiment for a Nyquist folded bandpass receiver adding a narrow bandfilter for enhancing the reception of ultra wideband (UWB) pulses isdescribed with respect to FIG. 8.

The reconfigurable RF bandpass sampling receiver of FIG. 1 and theanalog interpolation filter embodiments of FIGS. 2A and 2B are alleffective receiver architectures. The reconfigurable direct RF bandpasssampling receiver of FIG. 1 is particularly advantageous for ELINT typesystems and also for receiving wideband signals, more generally. Forexample, this RF bandpass sampling receiver can be used to detectwideband signals, such as those having a bandwidth of about a fewhundred MHz to about 1 GHz or more with the bandwidth being about 5% to15% of the RF center frequency. The RF bandpass sampling receiver canalso have a relatively large tuning range (e.g., from less than 2 GHz to20 GHz or more). However, this RF bandpass sampling receiver will likelyhave a moderate dynamic range because, as discussed below, the ADC inthis embodiment will typically need to be clocked as fast as RFpre-sampler circuitry is clocked. In addition, the RF pre-sampler andthe ADC will often both have to run fast enough to avoid aliasingproblems.

In addition to being advantageous for ELINT type systems, theembodiments utilizing analog interpolation filters, as discussed withrespect to FIGS. 2A and 2B below, are also advantageous for receivers incommunication systems. In these embodiments, the analog interpolationfilter acts in operation as an anti-aliasing filter for the ADC. Thismeans that the bandwidth for the receiver can be extremely narrow, ifdesired. These embodiments, therefore, are particularly useful forcommunications receivers having narrow bandwidth and high dynamic rangerequirements. In addition, these embodiments are useful for receiverimplementations needing large tuning range requirements, such as isoften the case for some multi-mode communications systems, because theseembodiments can be more effective at lower frequencies and at higherfrequencies than the embodiment of FIG. 1.

FIGS. 3, 4 and 7 further depicted example embodiments of Nyquist foldedbandpass sampling receivers. These configurations are advantageous formonitoring very large bandwidths with fairly sparse signal environments,and they are particularly advantageous for radar warning receivers(RWRs), electronic support (ES) receivers, electronic support systems(ESS) receivers, and electronic support measures (ESM) receivers. TheNyquist folding bandpass receiver has a wider bandwidth/dynamic rangecombination than other receiver technologies and architectures. Also,the Nyquist folding bandpass receiver can provide signal copy (wheremany really wideband receivers cannot provide signal copy), which meansthat the Nyquist folding bandpass receiver is capable of being used as acommunications receiver for certain applications. As depicted, there isno anti-aliasing filter for RF sampling in the Nyquist folding bandpassreceivers; however, an anti-aliasing filter can be included for the ADC.

It is noted that the Nyquist folding bandpass receiver can havedifficulties seeing really short pulsed signals. The reception of suchpulsed signals, such as ultra wideband (UWB) pulsed signals, is improvedfor the Nyquist folding bandpass receiver with the configuration of FIG.8. This embodiment adds a narrow band (NB) filter in parallel with thewideband, multi-Nyquist zone filter to improve reception of shortpulses.

Now looking first to FIG. 1, a reconfigurable direct radio frequency(RF) bandpass sampling receiver is described that provides direct RFsampling of an input signal spectrum passed through a bandpass filterthat is tunable over a wide frequency range of interest and which issampled based on the bandwidth of the filter rather than the bandwidthof the total frequency range of interest. The bandwidth of the filtermay further be variable to provide for optimized search against avariety of signal bandwidths. The reconfigurable direct RF bandpasssampling receiver may be implemented with a single fixed clock forapplications where the signals of interest lie in adjacentnon-overlapping frequency channels. For applications requiring arbitrarytuning, the reconfigurable direct RF bandpass sampling receiver can alsouse a programmable or switched sampling clock to avoid the Nyquistsample boundaries that occur with bandpass sampling and thus providesampling of arbitrary frequencies. A high-speed analog pre-sampler maybe included to extend the maximum frequency range that can be sampled.In an electronic intelligence (ELINT) application, this reconfigurabledirect RF bandpass sampling receiver architecture provides a method forachieving a much smaller receiver form factor than previously achieved.This receiver architecture also provides a way to achieve better spurperformance with less phase noise by tuning the filter, instead oftuning the RF signal, and then by performing bandpass sampling (i.e.,sampling at a non-DC frequency). This receiver architecture alsoprovides the optional use of two or more clock sampling frequencies(i.e., FS1, FS2 . . . ) to allow adjustment of the Nyquist boundarieswhich occur at integer multiples of FS/2.

Although direct RF down-sampling receivers using optical technology havebeen used, these receivers convert a fixed section of RF bandwidth andare not tunable. Although direct RF bandpass sampling receivers havebeen proposed, no solution has been presented in prior art to allowdirect RF bandpass sampling with tuning over a wide frequency range ofinterest. Although sampling with switched bandpass filters to implementadjacent channel tuning in a Nyquist-sampled sigma delta architecturehas been used, this architecture is fundamentally limited in tuning tohigher RF frequencies because of the need for the modulation clock tomeet Nyquist sampling criteria for the maximum frequency; furthermorethis architecture does not provide arbitrary tuning. In addition, directRF receivers based on the use of analog high-speed pre-samplers havebeen built, although not in any reconfigurable architecture. Thereconfigurable direct RF bandpass sampling receiver architectureeffectively combines aspects of other architectures into a particularlyadvantageous solution. This receiver architecture is particularlyadvantageous for applications requiring a wide range of centerfrequencies extending to very high frequencies such as where signalfrequencies are in a range from 2 GHz or lower to 10 or 20 GHz andhigher. As such, the receiver architecture is advantageous for us inwide frequency range of interest ELINT receiver implementations.

FIG. 1 is a block diagram of an embodiment 100 for a directradio-frequency (RF) bandpass sampling receiver. As depicted, the radiofrequency (RF) input signal 116 is first passed through an optionalpre-select bandpass filter (BPF) 118 and then to a low noise amplifier(LNA) 102. The output 130 of the LNA 102 is provided to a tunablebandpass filter 104, which can be configured to have a tunable centerfrequency and a programmable bandwidth dependent upon one or more filtercontrol signals 105. The filtered output signal 132 is received bysampler or sampling circuitry 106, which can include a high speed analogpre-sampler 108 and analog-to-digital converter (ADC) circuitry 110. Thedigital output signals from sampling circuitry 106 are then furtherprocessed by digital signal processing (DSP) circuitry 114 to producebaseband in-phase path (I) and quadrature path (Q) signals. The sampler106 receives a sampling clock (FS) input signal 121 at a desiredsampling frequency. This sampling clock (FS) signal 121 determines thesampling frequency for the sampling circuitry 106.

This sampling clock (FS) input signal 121 can be generated bymulti-clock generation circuitry 112. As depicted, the multi-clockgeneration circuitry 112 generates two or more clock signals that can beselected and used as sampling clocks (CLK1, CLK2 . . . ) for thesampling clock signal 121 that is provided to sampling circuitry 106. Asshown, two or more clock signals can be generated by the multi-clockgeneration circuitry 112, namely, a first clock signal (CLK1) 120, asecond clock signal (CLK2) 122, and any number of additional clocksignals, as desired. A clock select signal 126 is provided to themulti-clock generation circuitry 112 to select which of the clocksignals will be used as the output clock signal for the sampling clocksignal (FS) 121 to the sampling circuitry 106. It is also noted thatmulti-clock generation circuitry 112 could be implemented using a widevariety of clock circuits. For example, the multi-clock generationcircuitry 112 could be configured to always generate multiple clocksfrom which a sampling clock is selected. Alternatively, the multi-clockgeneration circuitry 112 could be configured to generate a single outputclock signal that is adjusted according to the clock select signal 126to provide a programmable clock output signal at the desired samplingfrequency. Other variations could be provided, as desired. It is alsonoted that in some embodiments a single clock signal could be utilized,if desired. In such an implementation, the clock generation circuitry112 would provide a single fixed sampling clock output signal for thesampling circuitry 106.

As described herein, it is noted that these different clock signalsallow for selection of an appropriate sampling clock based on meetingthe Nyquist criteria of the bandpass filter and based on criteria so asto avoid Nyquist sampling problems due to Nyquist boundaries. Nyquistzones are determined by the sampling rate for the sampling circuitry106, and Nyquist criteria locate sampling zone boundaries at integer(K=0, 1, 2, 3 . . . ) multiples of f_(S)/2 starting at DC (frequency=0Hz). In addition, Nyquist zones alternate between non-inverted andinverted spectrums. Traditional Nyquist criteria states that forbandpass sampling, the sampling rate must be two-times or greater thanthe bandwidth of the signal frequency range of interest, and that forbaseband sampling, the sampling rate must be two-times or greater thanthe maximum frequency for the signal frequency range of interest.

As frequency ranges within the signal input spectrum are analyzed,depending upon the sampling frequency for the sampling circuitry 106,one or more Nyquist boundaries could be crossed during processing. Thus,by having multiple sampling clock signals available, when a Nyquistboundary for a first sampling clock signal is being reached duringprocessing across a frequency range, a switch can be made to using asecond sampling clock signal. As such, the Nyquist boundary will alsochange based upon this new sampling frequency. In operation, therefore,if the tunable bandpass filter 104 is tuned to a new frequency and itsbandwidth includes a Nyquist boundary, a switch could be made to analternative sampling frequency so that reconstruction problems at theNyquist boundaries can be avoided. Thus, with proper selection of thesampling clock signals, the respective Nyquist zone boundaries for thesesampling clock frequencies can be made far enough apart so that Nyquistsampling problems for the first sampling clock can be avoided byswitching to the second sampling clock, and vice versa. In addition, asindicated above, the present invention is not limited to two clocksignals, and any number of selectable clock signal frequencies could beprovided, as desired.

The reconfigurable direct RF bandpass sampling receiver architecture hasa number of advantages. As a direct RF receiver architecture, thereceiver 100 can receive high frequency signals without using a mixer todown-convert the incoming RF signal. Instead, the receiver 100 reliesupon aliased bandpass sampling within the sampling circuitry 106 inconjunction with the tunable filter 104 to directly down-convert thereceived signals. To allow sampling of very high frequency RF signals, ahigh-speed analog pre-sampler may be used. Because this receiver doesnot utilize a mixer, it avoids phase noise, spurs and mixer productsthat typically result from generating a local oscillator (LO) mixingsignal and mixing it with the incoming RF signal. This receiver can alsohave faster response times in changing frequency because it does notneed the LO to settle. In addition, the receiver uses less hardware andallows for a smaller form factor because no mixer and no tuner areutilized. It is also noted that the clock performance for the clockgeneration circuitry is preferably tight in order to avoid ADC jittererrors. Again, the multiple clock signals that can be utilized for thesampling clock signal (FS) 121 are provided to help avoidnon-recoverable zones around Nyquist boundaries as the tunable filter104 is adjusted across a frequency band of interest for incoming signals116. Because tunable bandpass filters are used, the reconfigurabledirect RF bandpass sampling receiver can receive frequencies across awide range of frequencies, unlike previously used optical-based directRF down-sampling receivers. When implemented with multiple clocks, thereconfigurable direct RF bandpass sampling receiver can furthermore tuneto arbitrary frequencies across a wide range of frequencies.

Example circuitry that may be utilized for the sampling circuitry 106includes pre-samplers available from PICOSECOND PULSE LABS (PSPL) toallow sampling by an ADC in cases when the frequency of the signal isabove the effective bandwidth of the ADC. Example circuitry that may beutilized for the tunable filter 104 includes tunable filter banksavailable from PARATEK. If desired, other tunable filter technologiescould be utilized, such as tunable optical Mach-Zehnder filtertechnology, tunable image rejection notch filters, tunable bandpassfilters based on active inductor technology, tunable filter that usethin film ferroelectric varactors to provide voltage controlled phaseshifting, and tunable filters the use RF microelectromechanical systems(MEMS) technology.

Looking back to FIG. 1, it is noted that the preselect filter 118 isoptional component (depending on application) to suppress frequenciesoutside the range of interest. It is also noted that the LNA 102 can beused to increase the gain to within range of the sampling circuitry 106given any insertion losses of the tunable filter 104, and the LNA 102can be used to provide for a desired noise figure response. The LNA 102can be implemented as desired, and the specific implementation for theLNA 102 will be technology and application dependent. In addition, thefunctionality of the LNA 102 may be split between different component,as desired (e.g., the gain may be spread to optimize for a specificcascade analysis). In some strong signal applications, an LNA may not berequired.

The tunable filter 104 can be implemented as a narrowband filter thatcan be tuned over the frequency range of interest. For ELINTapplications, the low end of the frequency range might be from 500 MHzto 2 GHz and the high end of the frequency range might be over 20 GHz. Atypical instantaneous bandwidth of the tunable filter could be anywherefrom 5% to 15% or more of the center frequency, or the tunable filtercould be configured to have an instantaneous bandwidth ranging from 25MHz to over 1 GHz. The tuning can be implemented via continuous tuning(e.g., a voltage controlled dielectric) or by closely spaced switchedfilter banks, or by a combination of continuous tuning and switchedfilter banks. Other implementations could also be used, if desired.Regardless of how the tuning is implemented, the basic form of thereconfigurable direct RF bandpass sampling receiver is characterized bythe ability to perform bandpass sampling (i.e., sampling at non-DCfrequencies) with arbitrary center frequency over a large RF range ofcenter frequencies. It is noted that if the instantaneous bandwidthapproaches a large fraction of the center RF frequency, then thebenefits of the reconfigurable direct RF bandpass sampling receiver overfull bandwidth sampling will likely decrease. In a possibly limitingcase, the instantaneous bandwidth equals the full range of the frequencyrange of interest, and no tuning is required for the tunable filter 104.At this point, the reconfigurable direct RF bandpass sampling receiverbecomes similar to a direct RF bandpass sampling receiver.

The sampler 106 can be a module that inputs the band-limited RF signaland performs analog to digital conversion. The sampler module mayinclude a high-speed analog pre-sampler 108 configured to capture thesignal prior to being processed by a sample-and-hold circuit of atypical analog to digital converter (ADC) to allow sampling by an ADC incases when the frequency of the signal is above the effective bandwidthof the ADC. The ADC 110 can follow the pre-sampler 108. It is noted thatthe sampler module may be fully integrated and that for someapplications the pre-sampler may not be required.

The DSP 114 can include additional digital processing for the receivedsignals. For example, the DSP 114 can include filtering, decimation,conversion of the real sampled data to in-phase/quadrature-phase (I/Q)data, equalization, signal detection, and signal measurement. Additionalpost-processing DSP functions required by the specific application canalso be integrated into the DSP 114, as desired, for the reconfigurabledirect RF bandpass sampling receiver.

It is noted that there are key differences between all prior bandpasssampling techniques and the reconfigurable direct RF bandpass samplingreceiver. One important difference is that the receiver architectureuses a tunable bandpass filter 104 that can tune to many differentNyquist zones. A second important difference is that the receiverarchitecture can be tuned to any arbitrary frequency between the minimumand maximum frequency range. To help achieve this result while coveringmultiple Nyquist zones, the receiver architecture utilizes a secondclock. By using a second clock, the reconfigurable direct RF bandpasssampling receiver shifts the Nyquist boundary to a different location,allowing recovery of signals on the Nyquist boundary of the first clock.Prior solutions do not allow a continuous range of frequency coveragethat crosses multiple Nyquist boundaries.

The reconfigurable direct RF bandpass sampling receiver allows theflexibility of a reconfigurable heterodyne receiver with widebandfrequency coverage without any oscillator-based tuning hardware, thusreducing cost, size, weight, and power (CSWAP) and improving performancein spur performance, tuning speed, and phase noise. In addition, thereconfigurable direct RF bandpass sampling receiver has better dynamicrange and improved CSWAP than receivers with instantaneous bandwidthequal to the full frequency range (such as wideband compressivereceivers and full bandwidth digitizers). For typical wirelessapplications with frequency range under a few GHz, the reconfigurabledirect RF bandpass sampling receiver has similar advantages as a RFnoise shaping sigma delta digital receiver. However, as the maximumfrequency of interest increases, the reconfigurable direct RF bandpasssampling receiver is advantageous over the RF noise shaping sigma deltadigital receiver because the modulator sampling clock of the RF noiseshaping digital receiver is constrained by the Nyquist criteria of themaximum frequency and thus must be at least twice the maximum frequency.Thus for applications such as ELINT/EW/ESM, the reconfigurable direct RFbandpass sampling receiver is, therefore, superior to the RF noiseshaping sigma delta digital receiver.

It is noted that the reconfigurable direct RF bandpass sampling receiveralso uses bandpass sampling, and there are a few key differences betweenall of prior bandpass sampling receivers and the reconfigurable directRF bandpass sampling receiver. One important difference is that thereconfigurable direct RF bandpass sampling receiver uses a tunablebandpass filter that can tune to many different Nyquist zones. A secondimportant difference is that the reconfigurable direct RF bandpasssampling receiver can be tuned to any arbitrary frequency between theminimum and maximum frequency range. In order to achieve this resultwhile covering multiple Nyquist zones, the reconfigurable direct RFbandpass sampling receiver can use a second sampling clock so that thesignal can be reconstructed at the Nyquist boundaries. By using a secondclock, the reconfigurable direct RF bandpass sampling receivereffectively shifts the Nyquist boundary to a different location,allowing recovery of signals on the Nyquist boundary of the first clock.Prior solutions do not allow such a continuous range of frequencycoverage crossing multiple Nyquist boundaries. The multi-signal bandpasssampling concept is limited partially by the fact that the filters arenot reconfigurable, but more importantly, the filters are severelyconstrained in design to avoid overlapping aliased signal spectra. Thus,the reconfigurable direct RF bandpass sampling receiver more flexiblethan multi-signal bandpass sampling utilized in prior solutions.

Filter control signal 105 is typically used to set the tunable bandpassfilter to a particular frequency and bandwidth and is stationary for aperiod of time before being reconfigured. Alternatively, the filtercontrol signal 105 could be smoothly or rapidly changing (time-varying)in frequency and/or bandwidth similar to the swept bandpass filterreceiver described in more detail in U.S. patent application Ser. No.11/247,314 entitled “SWEPT BANDPASS FILTER FREQUENCY MODULATEDCONTINUOUS WAVE (FMCW) RECEIVER AND RELATED METHODS,” which was filed onOct. 11, 2005, and which is hereby expressly incorporated by referencein its entirety. In the reconfigurable direct RF bandpass samplingreceiver, however, the time-varying filter avoids having Nyquistboundaries inside the filter bandwidth. Nyquist boundaries may beavoided in the time-varying case by configuring the filter bandwidth tobe less than the bandwidth of the particular Nyquist zone (i.e.,bandpass region of interest) so that as the filter center frequencychanges, the filter bandwidth is always within a single Nyquist zone, orthey may be avoided by tuning to separate Nyquist zones while avoidingthe boundaries by use of appropriate center frequency and filterbandwidth, or they may be avoided by time-varying the clockappropriately, or through a combination of these techniques. It is notedthat in the case where the clock is time-varying to allow thetime-varying filter to avoid Nyquist boundaries, signal reconstructionmay be difficult, and thus a smoothly or rapidly changing (time-varying)clock would typically be avoided.

Because the physical architecture of the reconfigurable direct RFbandpass sampling receiver is similar to the physical architecture ofthe swept bandpass filter FMCW receiver, in principle the reconfigurabledirect RF bandpass sampling receiver could be modified to function as aswept bandpass filter FMCW receiver by allowing the filter to betime-varying and by allowing the filter to cross Nyquist boundaries andby suitable modification of the DSP. Similarly, the swept bandpassfilter FMCW receiver could, in principle, be modified to function as asingle-clock embodiment of the direct RF bandpass sampling receiver bydiscrete tuning the frequency/bandwidth (rather than time-varying) or byappropriate choice of clock so that Nyquist boundaries are avoided asthe filter sweeps, and by suitable modification of the DSP to allowreconstruction of the captured time-frequency bandwidth. It is notedthat because of typical component limitations, a combined functionalityreceiver for either direct RF bandpass sampling and/or swept bandpassFMCW receiver would not perform as well and/or would cost significantlymore; thus while a combined receiver could, in principle be built, thiswould not represent best practice. The swept bandpass FMCW receiverprovides the best benefit as an IF receiver with a wide-band front-endtuner reducing the RF frequency to the low GHz range with the DSPoptimized for detection and measurement of wide bandwidth FMCW signals,while the direct RF bandpass sampling receiver provides the best benefitas an RF receiver operating over a wide range of frequencies from thelow GHz range to 20 GHz and above with the DSP optimized for generalpurpose signal reconstruction, detection, and measurement.

Bandpass receiver architectures that utilize an analog interpolationfilter are now discussed with respect to FIGS. 2A and 2B. In particular,FIG. 2A provides a further embodiment where an analog interpolationfilter is added to the reconfigurable direct RF bandpass samplingreceiver discussed with respect to FIG. 1. And FIG. 2B is a Nyquist zonediagram associated with architecture of FIG. 2A.

As discussed above, the embodiment 100 of FIG. 1 provides for the use ofmultiple clocks by sampler 106 such that the operation of thepre-sampler 108 and the ADC 110 can avoid Nyquist zone boundaries. Inpart, the embodiment 100 provides a solution that allows for a muchsmaller form factor for applications requiring extremely wide tuningbandwidth. It also provides a way to achieve better spur performance bytuning the filter instead of timing the RF signal and then performingbandpass sampling with a second clock to allow adjustment of the Nyquistboundaries (which occur at integer multiples of F_(S)/2). One potentialproblem with the embodiment 100 of FIG. 1, however, is that thepre-sampler 108 can act as a type of extended sample-and-hold for theanalog-to-digital converter (ADC) 110 so that the quantization rate ofthe ADC 110 will equal the RF bandpass sample rate of the pre-sampler108. This implies that the digital signal processing (DSP) 114 will beinteracting with a switchable or tunable sampling clock signal (F_(S))121. In addition, it may be difficult to use a slower-speedhigh-dynamic-range ADC because the RF bandpass sample rate willtypically be high. A high RF bandpass sample rate is preferred ingeneral because the dynamic range of the RF bandpass sampling is greaterand because this allows a wider transition region for the tunable orswitchable filter 104 thereby simplifying the filter design. Anotherproblem caused by the requirement for synchronous sampling is that inimplementations where the pre-sampler 108 and ADC 110 are separatephysical devices (which is typically the case for sampling for highfrequency RF signals with current technology) is that precision delaysmust typically be used. For example, a delay can be added to a commonclock signal used by the pre-sampler 108 and the ADC 110 in order toprovide for synchronous sampling.

FIG. 2A provides an embodiment where an analog interpolation filter isadded to the reconfigurable direct RF bandpass sampling receiverdiscussed with respect to FIG. 1. The addition of the analoginterpolation filter allows the ADC clock to be de-coupled, andpotentially completely de-coupled, from the RF sample clock. Thus, thismodification of the architecture of FIG. 1 allows the quantization to beperformed at a much slower rate than the initial sampling and allows thefinal analog bandwidth to be much narrower than the bandwidth of thefirst stage filter located before the high-speed sampler. Thecombination of a tunable bandpass filter, tunable bandpass sample clockand analog interpolation filter followed by a further stage of samplingand quantization at a slower rate than the bandpass sample clockingprovides significant advantageous by de-coupling the quantization samplerate from the high-speed sample rate. As with the embodiment 100 of FIG.1, the embodiment 200 of FIG. 2 provides a simplified receiverarchitecture capable of covering extremely large RF ranges (e.g., fromless then 2 GHz to greater than 20 GHz) and suitable for a variety ofsignal applications including electronic intelligence (ELINT) signalapplications with typical instantaneous bandwidth of 5% to 15% of thecenter frequency. Further, the addition of the narrower analoginterpolation filter also allows arbitrarily narrow instantaneousbandwidth for various applications including communications andnarrow-band searching.

Looking now in more detail to FIG. 2A, a block diagram is depicted foran example embodiment 200 of a reconfigurable direct RF bandpasssampling receiver with an analog interpolation filter 206. The overallreceiver architecture for embodiment 200 is similar to that forembodiment 100 of FIG. 1. However, the functionality of the sampler 106of FIG. 1, which included pre-sampler 108 and ADC 110, has beenimplemented as a non-quantizing sampler 204, an analog interpolationfilter 206, intermediate frequency (IF) amplifier 208 and ADC 210. Thenon-quantizing sampler 204 receives a RF sampling clock signal from RFsample clock circuitry 212, and the ADC 210 receives a quantizationsampling clock signal from ADC sample clock circuitry 214. Because theADC 210 has been de-coupled from the non-quantizing sampler 204 throughthe analog interpolation filter 206, the RF sample clock signal and theADC sample clock signal can be different. It is noted that RF sampleclock circuitry 212 can also be configured to provide a tunable orselectable RF sample clock, if desired. As discussed above with respectto clock circuitry 112 in FIG. 1, the RF sample clock circuitry 212 cangenerate a clock signal having a tunable frequency or can generate anumber of selectable clock signals.

As further depicted in FIG. 2, the antenna 202 provides an RF inputsignal to the RF low noise amplifier (RF LNA) 102. This amplified signalis then provided to the tunable bandpass filter 104, which is discussedabove. The signal is then processed by the non-quantizing sampler 204,passed through the analog interpolation filter 206, and then digitizedwith ADC 210. As shown, an additional amplifier 208 can be providedbefore the ADC 210 which amplifies the output of the analoginterpolation filter 206 and is configured to operate at theintermediate frequency (IF) output by the analog interpolation filter206.

Numerous advantages are achieved through the addition of the analoginterpolation filter 206 between the high-speed non-quantizing sampler204 and the ADC 210. This implementation allows the quantization samplerate of the ADC 210 to be de-coupled, and potentially completelyde-coupled, from the RF bandpass sample rate of the non-quantizingsampler 204. In particular, in comparison with embodiment 100 of FIG. 1,the embodiment 200 of FIG. 2 allows the quantization in ADC 210 to occurat a much slower rate than the rate of the ADC 110 and sampler 108within the sampler 106 of FIG. 1.

It is noted that the analog interpolation filter 206 is a filter havingbandwidth less than or equal to the Nyquist bandwidth of the high-speednon-quantizing sampler 204 in order to help provide smoothinterpolation. This interpolation greatly facilitates the de-coupling ofthe high-speed RF sample rate of the non-quantizing sampler 204 from thequantization sample rate of the ADC 210. It is also noted that theanalog interpolation filter 206 is different in function from a widebandIF (intermediate frequency) filter that could be placed at the output ofa high-speed sampler 106 in FIG. 1 in order to help stretch the pulse aspart of a sample-and-hold process. Such a wideband IF filter forpurposes of stretching the pulse has been used, for example, withrespect to pre-samplers available from PICOSECOND PULSE LABS (PSPL).

FIG. 2B is a Nyquist zone diagram related to the operation of the analoginterpolation filter. As depicted, the diagram 250 represents thesampled signal versus frequency and shows the Nyquist zones createdbased upon the RF sampling rate (F_(S)) from the RF sample clock 212,which can be a tunable or selectable RF sample clock as discussed above.In particular, a baseband Nyquist zone (NZ0) is created between zero andF_(S)/2. A first Nyquist zone image (NZ1) is created between F_(S)/2 andF_(S). A second Nyquist zone image (NZ2) is created between F_(S) and3F_(S)/2, and so on. The quantization sample clock for the ADC 210 canthen be operated to determine the Nyquist zone from which the signalswill be quantized. The filter response for the analog interpolationfilter 206 is represented as filter response (F) 252 in the basebandNyquist zone (NZ0). The filter response then also imaged into eachNyquist zone image as represented by arrows 262, 264 and 266. As shown,the analog interpolation filter 206 is seen as filter response (F) 254in the first Nyquist zone image (NZ1), as filter response (F) 256 in thesecond Nyquist zone image (NZ2) 256, and so on.

With respect to the quantization sample clock for ADC 210, thequantization sample clock could be made to be fixed for applications inwhich the desired frequency range can be received without tuning thequantization sample clock. If desired, the quantization clock for theADC 210 could also be implemented as a tunable clock signal to providegreater flexibility in overall system tuning to frequencies of interest.For example, the quantization clock would typically be implemented as atunable clock if the ADC 210 is performing bandpass sampling and if theinterpolation filter is tunable in order to properly perform bandpasssampling. On the other hand, if the interpolation filter is tunable in ahybrid architecture with an RF tuning element after the interpolationfilter, then the quantization clock sample rate would typically bedetermined by the bandwidth requirements and not the bandpass samplingrequirements.

In operation, the embodiment 200 of FIG. 2A provides the advantage ofde-coupling the high-speed sampling and quantization processes by usingan analog interpolation filter after the bandpass sampling and beforethe quantization. As such, the sampling and quantization can becompletely de-coupled from the RF bandpass sampling process, if desired,and the Nyquist zone within which the ADC 210 operates can be selectedas shown with respect to FIG. 2B. In prior bandpass sampling, thesampling/quantization process is typically performed at the same time asthe RF sampling, with most of the power requirements tied to thequantization function rather than the RF sampling function. The analoginterpolation filter provides significant advantages includingsimplifying the ADC timing, simplifying the DSP processing of thequantized output signals, reducing total power consumption, allowing fora high-dynamic range ADC to be used, allowing for tunable bandpasssampling at an intermediate frequency (IF), and allowing for multi-stagebandpass sampling.

As discussed above, the reconfigurable direct RF bandpass samplingreceiver of FIG. 1 provides a method for achieving a smaller form factorthan previously possible for applications requiring extremely widetuning bandwidth and also provides a way to achieve better spurperformance by tuning the filter instead of tuning the RF signal andthen performing bandpass sampling with tunable and/or multiple clocksignals to allow adjustment of the Nyquist boundaries. The addition ofan analog interpolation filter, as shown with respect to FIG. 2, allowsthe quantization sample rate to be de-coupled from the RF bandpasssample rate.

Nyquist folded bandpass sampling receivers, which can utilize thereceiver architectures described with respect to FIG. 1 and FIG. 2Aabove, are now discussed with respect to FIGS. 3, 4, 5A, 5B, 6A, 6B, 7and 8. In particular, FIG. 3 depicts an embodiment of a Nyquist foldedbandpass sampling receiver that uses an ultra wideband (UWB) front-endfilter so that multiple Nyquist zones are received. Thus, for thisNyquist folded bandpass sampling receiver, the filter 302 serves as apre-select filter instead of serving as an anti-aliasing filter for theRF-sampler 204. FIG. 4 depicts a further embodiment related to FIG. 3.FIGS. 5A, 5B, 6A and 6B provide signal diagrams related to the operationof the Nyquist folded bandpass sampling receiver. FIG. 7 provides aNyquist folded bandpass sampling receiver that does not use an analoginterpolation filter to de-couple the RF sampling clock and thequantization clock. Although the de-coupling of the RF bandpass sampleclock and the quantization clock is helpful for the Nyquist foldedbandpass sampling receiver, having separate clocks is not required, andthe present invention includes implementations using other clockarrangements such as where the clocks are synchronized, for example,with the embodiment of FIG. 7. FIG. 8 provides a Nyquist folded bandpasssampling receiver including the addition of a narrow band filter in thefront end to improve the reception of UWB pulsed signals because,without an anti-aliasing filter, UWB pulses are not stretched by theanti-aliasing filter and are likely to pass through the RF samplerwithout being seen.

FIG. 3 is a block diagram for an example embodiment 300 of a direct RFbandpass sampling receiver with a ultra wideband (UWB) front end filterto allow reception of multiple Nyquist zones. Similar to embodiment 200of FIG. 2A, embodiment 300 includes a non-quantizing RF sampler 204followed by an analog interpolation filter 206, an intermediatefrequency amplifier (IF Amp) 208, and an analog to digital converter(ADC) 210. As with embodiment 200, the ADC 210 receives an ADC samplingclock signal from ADC clock circuitry 214. In addition, a digital signalprocessor (DSP) may receive digital signals from the ADC 210 and processthese digital signals to identify and remove the Nyquist-zone specificmodulation in order to recover the original signal. Thus, the DSPcircuitry can be configured to remove the induced modulation associatedwith the Nyquist zone in which the signal was located and to acquire theoriginal signal by utilizing the Nyquist zone in which the signal waslocated, removing an induced modulation for that Nyquist zone, andacquiring the original signal.

In contrast with embodiment 200 of FIG. 2A, embodiment 300 includes anultra wideband (UWB) filter 302 in front of the non-quantizing RFsampler 204 and uses modulated RF sample clock circuitry 304. Thewideband filter 302 has a bandwidth that is wide enough to pass multipleNyquist zones where the Nyquist zones are determined by the RF samplingclock frequency for the non-quantizing RF sampler 204. The modulatedsample clock circuitry 304 provides an RF sampling clock signal to thenon-quantizing RF sampler 204 that is not constant and is adjusted ormodulated during sampling. For example, the modulation could be achirped sampling clock signal, although other types of modulated clocksignals could also be used depending upon the results desired. In oneexample implementation, the wideband filter circuitry can be configuredto have a bandwidth of 20 GHz or more, and the modulated sampling clocksignal can be generated to have a sampling rate of 2 GHz or more. Otherimplementations could be made, as desired.

FIG. 4 is a block diagram for an embodiment 400 for a Nyquist foldedbandpass sampling receiver using a switchable chirped sampling clocksignal and a fixed low-pass interpolation filter. As depicted, theembodiment 400 has a switchable modulated sample clock for the RF sampleclock 304. The fixed interpolation filter is implemented as a low-passfilter with a 3-dB bandwidth of about 1 GHz and a stop-band bandwidth ofabout 1.25 GHz meeting the anti-aliasing requirements of the ADC 210. Inaddition, the embodiment 400 in FIG. 4 includes a pre-select filterpassing frequencies from 1 GHz to 20 GHz as the front-end widebandfilter 302. Also as depicted, the switchable modulated sample clockcircuitry 304 provides a linear chirp clock signal that has a bandwidthof about 10 MHz and can be configured to have two RF carriers centeredat about 1.9 GHz and at about 1.8 GHz. Further, the ADC 210 has aquantization sampling rate of about 2.5 Giga-samples per second (Gsps).The receiver depicted in FIG. 4 has an instantaneous bandwidth (IBW) ofover 15 GHz. A total of about 20 Nyquist zones with bandwidth about 950MHz and about 800 MHz (depending on the center frequency of themodulated RF sample clock 304) are seen by the RF sampler 204, and theinterpolation filter 206 passes all of each Nyquist zone. The ADC 210,therefore, sees over 15 GHz folded into the interpolation filter'sinstantaneous bandwidth of roughly 1 GHz. Despite this extremely largeIBW, however, the example Nyquist folding receiver of FIG. 4 will likelynot be able to process accurately pulses with duration less than 500pico-seconds since extremely short pulses may fall between samples inthe RF sampler for this particular RF sample rate. The embodiment inFIG. 8, as discussed below, can be used to address improved reception ofshort pulses.

The bandpass sampling Nyquist folding receivers, therefore, uses Nyquistfolding in conjunction with non-uniform sampling to enable extremelywide bandwidth receivers. In operation, multiple Nyquist zones arepassed through the wideband filter 302, and these Nyquist zones areallowed to fold on top of each other during sampling. Because the RFsampling clock is modulated, however, the signals that are foldedtogether from different Nyquist zones can be identified anddistinguished. For example, by using a shallow triangle frequency chirpas depicted in FIG. 4, separate modulations are induced within eachNyquist zone. As such, when the Nyquist zones fold on top of each other,the different signals from different Nyquist zones can be separated andidentified based on the fact that the induced modulation is differentfor each Nyquist zone. Thus, by using one or more clock modulations toinduce frequency modulations that are Nyquist zone dependent, multipleNyquist zones can be aliased together while still allowing for signalsfrom different Nyquist zones to be separated and identified. Because theinduced modulations can be measured, a determination can be made withoutambiguity of the Nyquist zone from which the signal originated. Thewideband filter and modulated sampling clock, therefore, make itpossible to cover an extremely wide bandwidth with a relatively slow,high dynamic range ADC in environments where the signal density isrelatively sparse. It is noted that the wideband filter, if desired, canhave a bandwidth of 10 GHz or 20 GHz or more and the associated samplingrate can be 1 GHz or 2 GHz or more. Other wideband filters could also beutilized, and the bandwidth could be selected as desired depending uponthe rate of the sampling clock and the number of Nyquist zones desiredto be passed It is also noted that the wideband pre-select filter 302does not need to be contiguous. Thus, different sub-regions of the totalfrequency span of interest may be selected simultaneously without regardto aliasing issues instead of selecting the total frequency span. It isalso noted that the pre-select filter 302 may be tunable and orreconfigurable in other ways. For example, in the case where thepre-select filter 302 selects sub-regions, it may be advantageous totune the sub-regions to cover different areas in response to jammers orto time-varying frequency regions of interest.

It is noted that the modulated clock signal can be configured as desiredto generate non-uniform sampling. For example, as used above, a chirpfrequency modulated RF sampling clock signal could be used, and thechirp could be based upon a linear sawtooth modulation (sometimesreferred to as a linear chirp), a sinusoidal modulation (sometimesreferred to as sinusoidal frequency chirp), a triangle frequencymodulation (sometimes called a triangle chirp), and/or any other desiredchirp modulation scheme. Other modulated clock signals and combinationsof modulations could also be used to provide non-uniform sampling,including frequency shift key, frequency agile, phase shift, generalfrequency modulation, etc. One combination that could be used is atriangle frequency modulation combined with an FSK (frequency shift key)modulation. It is also noted that the clock modulation may be switchableas described above, or it may be otherwise reconfigurable. For example,it may be advantageous to change the clock modulation from one type ofmodulation to another type of modulation in order to improve theperformance against different classes of signals.

As discussed here, this non-uniform sampling allows for the frequencymodulations generated by sampling to be different in different Nyquistzones. As long as a modulated sampling clock is used that will generatethis result, then the desired advantages of the Nyquist folded bandpasssampling receiver architectures described herein can be achieved. Assuch, a single clock modulation, or multiple clock modulations,mathematically translate into different signal modulations dependingupon the Nyquist zone in which the signals are located before beingfolded together thereby allowing separation of the aliased signals anddetermination of the Nyquist zone from which they came. It is furthernoted that the modulated sampling clock may be tunable or switchablesuch that the frequency of the clock signal may be tuned to a desiredfrequency and/or one of a plurality of generated clock signals may beselected. In addition, the modulated sampling clock may be configuredsuch that the modulation for the modulated sampling clock is adjustedduring operation of the receiver. Other variations and implementationscould also be utilized, if desired.

For example, a shallow linear chirp can be used for the RF samplingclock to generate non-uniform sampling. Assuming the shallow linearchirp varies the sampling clock according to a slope X, signals in theNth Nyquist zone will have induced slope magnitude given byX*ceiling(N/2) when they are uniformly re-sampled after theinterpolation filter, and the slope direction will be alternating foradjacent Nyquist zones because of spectral reversal. Assuming, forexample, the wideband filter 302 allows an extremely wide inputbandwidth between DC (0 GHz) to about 20 GHz and sampling occurs at 2Giga-samples per second (Gsps) for a plurality of relatively sparsepulsed signals, twenty Nyquist zones will be aliased into each otherduring sampling, and a measurement of the chirp slope on each pulse canbe used to determine from which Nyquist zone each pulse came. Thebaseband Nyquist zone covering DC to about 1 GHz will have no inducedslope. The first Nyquist zone covering about 1 GHz to about 2 GHz willhave induced slope=X. The second Nyquist zone covering about 2 GHz toabout 3 GHz will have induced slope=−X. The third Nyquist zone coveringabout 3 GHz to about 4 GHz will have induced slope=2X, and so on. Theability to make this determination, however, will typically rely uponthe pulse signal density being fairly low. It is noted that the signalinputs are not restricted to pulses and that pulsed signals are used inthis example to simplify the discussion. A more specific example using alinear chirp modulation for the sampling clock is now described withrespect to FIGS. 5A and 5B.

FIG. 5A is signal diagram 500 for a modulated RF sampling clock usingchirped or ramped clock signal. As depicted, the RF sample clockfrequency ramps between a rate of 2000 Mega-samples per second (Msps)and 2010 Msps about every 12 micro-seconds (μsec). In particular, the RFsample clock signal starts at about 2000 Msps and ramps to about 2010Msps from 0 to 12 μsec as represented by segment 502. The sample clocksignal then drops back down to 2000 Msps as represented by segment 510.This modulation of the sample clock signal then repeats as representedby ramping segments 504, 506 and 508 and by reset segments 512 and 514.As such, the RF sampling clock signal received by the non-quantizing RFsampler 204 is not constant and is adjusted as the sampling occurs.

FIG. 5B is signal diagram 550 for instantaneous frequency of tworeceived signals using the modulated RF sampling clock of FIG. 5A afterpassing through an interpolation filter located in the baseband Nyquistzone. For this example, the first received signal is at 2100 MHz and hasa pulse width of 1 μsec and a pulse repetition interval of 4 μsec. Thesecond signal is at 10120 MHz and has a pulse width of 1 μsec and apulse repetition interval of 6 μsec. As shown in FIG. 5A, the RFsampling clock is a linear sawtooth that ramps from 2000 Msps to 2010Msps with a period of 12 μsec. For this example, the first signal islocated in the second Nyquist zone (ie., from 2000 MHz (f_(S)) to 3000MHz (3f_(S)/2) when counting the baseband Nyquist zone from 0 MHz(f_(S)) to 1000 MHz (f_(S)/2) as the zero^(th) Nyquist zone). The secondsignal is located in the tenth Nyquist zone from (i.e., from 10000 MHz(5f_(S)) to 11000 MHz (11f_(S)/2)).

Looking to FIG. 5B, the instantaneous frequency for the first receivedsignal is represented by a first set of samples 552 that are obtainedduring the first ramp segment 502 in FIG. 5A. Instantaneous frequencyfor the second received signal is represented by a second set of samples554 that are obtained during the first ramp segment 502 in FIG. 5A.During the second ramp segment 504 in FIG. 5A, another set of samples562 are obtained that are related to the first received signal, andanother set of samples 564 are obtained that are related to the secondreceived signal. This repeats again for the third ramp segment 406 inFIG. 5A where a set of samples 572 are obtained for the first receivedsignal and where a set of samples 574 are obtained for the secondreceived signal. Only a portion of the time for the next ramp segment508 in FIG. 5A is depicted, where samples 582 and 584 are obtained. Fromthe sets of samples 552, 562 and 572 for the first received signals andthe sets of samples 554, 564 and 574 for the second received signals, itcan be discerned that the second signal's frequency slope issignificantly steeper than the first signal's frequency slope. As such,the slope can be measured, and the original Nyquist zone from whichthese signals originated can determined even though these two signalswill have been folded together during sampling. It is also noted thatsignals from odd-numbered Nyquist zones (1st, 3rd, etc.) will haveopposite induced modulation slope than signals from even-numberedNyquist zones.

Thus, by modulating the RF sampling clock signal to produce non-uniformsampling, such as through the use of the linear chirped signal in FIG.5A, the instantaneous frequencies obtained for different receivedsignals will appear with a particular signature, such as the differentslops related to each sample set 552, 562 and 572 for the first signaland each sample set 554, 564 and 574 for the second signal, as depictedin FIG. 5B. Even though the Nyquist zones fold on top of each other,different signals from different Nyquist zones can be separated andidentified based on the fact that the added modulation is different foreach Nyquist zone. The use of one or more clock modulations to providenon-uniform sampling and to induce frequency modulations that areNyquist zone dependent allows multiple Nyquist zones can be aliasedtogether while still allowing for signals from different Nyquist zonesto be separated and identified. This Nyquist folding is furtherdescribed with respect to FIGS. 6A and 6B.

FIG. 6A is a Nyquist zone diagram 600 for three signals received whenthe modulated RF sampling clock signal is at a first frequency. In thisexample, the sampling clock signal (f_(S1)) is 2000 MHz. Thus, a firstsignal (S1) sitting at 2200 MHz is located in Nyquist zone 2 (i.e., from2000 MHz (f_(S)) to 3000 MHz (3f_(S)/2)) and is 200 MHz away from thesampling clock signal (f_(S1)). Other signals may be indistinguishablefrom this signal once sampled if they or their images are also 200 MHzfrom the sampling clock signal (f_(S1)) or a harmonic (2f_(S1), 3f_(S1),4f_(S1)) of the sampling clock signal (f_(S1)). For example, a secondsignal (S2) sitting at 4200 MHz is located in Nyquist zone 4 (i.e., from4000 MHz (2f_(S)) to 5000 MHz (5f_(S)/2)) and is 200 MHz away from thesecond harmonic (2f_(S1)) of the sampling clock signal (f_(S1)). Oncesampled, this second signal (S2) will be folded on top of the firstsignal S1 and may be difficult to distinguish from the first signal S1.Similarly, a third signal (S3) sitting at 6200 MHz is located in Nyquistzone 6 (i.e., from 6000 MHz (3f_(S)) to 7000 MHz (7f_(S)/2)) and is 200MHz away from the third harmonic (3f_(S1)) of the sampling clock signal(f_(S1)). Once sample, this third signal (S3) is also folded on top ofthe first signal S1 and may be difficult to distinguish from the firstsignal (S1) and/or the second signal (S2). It is seen, therefore, thatby allowing multiple Nyquist zones through the wideband filter 302,signals within these Nyquist zones will be folded onto each otherthrough the sampling process. However, as discussed above, by modulatingor adjusting the sampling clock signal, these signals can bedistinguished. This is further depicted with respect to FIG. 6B.

FIG. 6B is a Nyquist zone diagram 650 for the three signals received inFIG. 6A when the modulated RF sampling clock signal is at a secondfrequency. As depicted, the adjusted sampling clock signal (f_(S2)) is2010 MHz. Thus, the first signal (S1) sitting at 2200 MHz is 190 MHzaway from the adjusted sampling clock signal (f_(S2)). The other signals(S2 and S3) are now distinguishable from this signal because they are nolonger 200 MHz from the sampling clock signal (f_(S2)) or a harmonic(2f_(S2), 3f_(S2), 4f_(S2)) of the sampling clock signal (f_(S2)). Forexample, the second signal (S2) sitting at 4200 MHz is now 180 MHz awayfrom the second harmonic (2f_(S2)) of the sampling clock signal(f_(S2)). Similarly, the third signal (S3) sitting at 6200 MHz is now170 MHz away from the third harmonic (3f_(S2)) of the sampling clocksignal (f_(S2)). Thus, by modulating or adjusting the sampling clocksignal, the three signals can now be distinguished from each other.

It is noted that the Nyquist zones in FIGS. 6A and 6B are labeled basedupon the assumption that the baseband Nyquist zone from f_(S) to f_(S)/2is counted as the zero^(th) Nyquist zone. As such, the first Nyquistzone (NZ1) is from f_(S)/2 to f_(S) (not shown). The second Nyquist zone(NZ2) is from f_(S) to 3f_(S)/2. The third Nyquist zone (NZ3) is from3f_(S)/2 to 2f_(S). The fourth Nyquist zone (NZ4) is from 2f_(S) to5f_(S)/2. The fifth Nyquist zone (NZ5) is from 5f_(S)/2 to 3f_(S). Thesixth Nyquist zone (NZ6) is from 3f_(S) to 7f_(S)/2. The seventh Nyquistzone (NZ7) is from 7f_(S)/2 to 4f_(S), and so on. Signals in each ofthese Nyquist zones will be folded on top of each other during sampling,assuming that the wideband filter 302 allows these Nyquist zones to passthrough to the RF sampling circuitry.

FIG. 7 is a block diagram for an embodiment 700 a Nyquist foldedbandpass sampling receiver that does not use an analog interpolationfilter to de-couple the RF sampling clock and the quantization clock. Asdepicted and described above, a wideband filter 302 has a bandwidth thatis wide enough to cover multiple Nyquist zones. The modulated sampleclock circuitry 304 provides a modulated sampling clock signal to thesampler 106. As with the embodiment 100 of FIG. 1, the sampler 106includes a pre-sample 108 and an ADC 110. The output of the ADC 110 isprovided the digital signal processor (DSP) 114. It is further notedthat other variations and architectures could be utilized while stilltaking advantage of a wideband filter 304 and a modulated sampling clockcircuitry 304. As discussed above, the wideband filter has a bandwidthwide enough to allow multiple Nyquist zones (based upon the RF samplingclock signal) to pass through, and the modulated sampling clockcircuitry 304 provides a mechanism for distinguishing signals withindifferent Nyquist zones that end up being folded together during the RFsampling process.

One potential problem with the Nyquist folded bandpass sampling receiveris its ability to handle pulsed signals with very wide bandwidths suchas ultra wideband (UWB) pulsed signals. As indicated with respect toFIG. 4 above, the example Nyquist folding receiver of FIG. 4 will likelynot be able to process accurately pulses with duration less than 500picoseconds, or bandwidth greater than approximately about 2 GHz. Toovercome this potential problem, a narrow band filter can be added inparallel with the wideband filter 302 at the front end of the Nyquistfolded bandpass sampling receiver so that UWB pulses can be moreaccurately processed. FIG. 8 provides such an embodiment.

FIG. 8 is a block diagram for an embodiment 800 of a Nyquist foldedbandpass sampling receiver having a narrow band (NB) filter 802 toimprove reception of ultra wideband signals. In particular, the RF inputsignal is provided both to the ultra wideband (UWB) filter 302, which iswide enough to cover multiple Nyquist zones as discussed above, and tothe narrow band (NB) filter 802. The outputs of the wideband filter 302and the narrow band filter 802 are then combined before being sampled bythe non-quantizing RF sampler 204. The rest of the embodiment 800 issimilar to the embodiment 300 of FIG. 3 as discussed above. As with theother embodiments described herein, the intermediate frequency amplifier(IF Amp) 208 can be located where desired and is not required to beplaced in front of the ADC 210. If desired, there may be a low-noiseamplifier (LNA) at any point in the receiver path.

The addition of the narrow band filter 802 facilitates the reception ofultra wideband signal pulses and, therefore, extends the Nyquist foldingbandpass sampling receiver to allow processing of ultra wideband (UWB)pulses. The UWB pulses that enter the wideband filter 302 will passthrough essentially unmodified. Thus, with a high bandwidth RF samplersuitable for direct RF sampling, the sample pulses are relatively narrowand the probability of a UWB pulse being completely missed by the RFsampler can be relatively high. However, any UWB pulse entering thenarrow-band filter 802 will be stretched in the time domain. Thus, theRF sampler can capture the UWB pulse particularly if the bandwidth ofthe narrow-band filter 802 is less than the Nyquist bandwidth of the RFsample rate. It is noted that the bandwidth of the narrow-band filter802 could be designed to be wider than the Nyquist bandwidth associatedwith rate of the RF sampling clock; however, the effectiveness of thisnarrow-band filter 802 for receiving UWB pulses would likely bedegraded.

The UWB pulses can be processed using any desired technique. Forexample, the UWB pulses can then be processed using known processingtechniques, such as FRI (Finite Rate of Innovation) techniques, with themodification to the processing to take into account the modulated sampleclock. Alternatively, individual UWB pulses can be processed by lookingfor the impulse response of the narrow-band filter, again taking intoaccount the modulated sample clock and interpolation filter followed byADC. Still further, the UWB pulses may be processed using knownfrequency domain processing techniques, such as a modified root MUSICtechnique applied to frequency domain information. It is further notedthat the narrow-band filter may be designed based on the sampling rateof the RF sampler sample rate, or it may be designed based on the FRItechniques.

It is noted that the parallel narrow-band filter 802 may be tunable, ifdesired. If tunable, the architecture has the advantage of allowingplacement of the narrow band filter 802 in a region of the RF range ofinterest that is relatively free from noise and interferers. Thisselected placement can advantageously provide for higher signal to noiseratio (SNR), provide for higher signal to interference ratio (SIR),and/or, more generally, provide for higher signal to interference andnoise ratio (SINR). Still further, multiple parallel narrow-band filtersmay be used, if desired. As such, additional filters would be coupledbetween the RF input signal and the RF sampling circuitry in parallelwith the wideband filter 302. If these additional narrow band filterswere configured to have the same impulse response as the first narrowband filter 802, then the impulse responses will coherently add therebyallowing for higher SINR. Thus, if desired, the narrow band filtercircuitry can be implemented as one or more tunable narrow band filterseach having a tunable center frequency dependent upon a filter controlsignal and each being tunable across the frequency range of interest.

With respect FIGS. 9A and 9B, an embodiment is now discussed for aNyquist folded bandpass sampling receiver architecture that combinesfeatures from FIG. 1, FIG. 2A and FIG. 3 to form a hybrid receiverarchitecture.

As described above, the reconfigurable direct RF bandpass samplingreceiver of FIG. 1 and FIG. 2A, can use a tunable and/or switchableanti-aliasing filter in conjunction with a tunable and/or switchable RFsampling clock in order to perform reconfigurable bandpass sampling downconversion. Although the analog interpolation filter is not required, itcan simplify the hardware design and provide more flexibility in designbecause it allows the RF sample rate to be completely de-coupled fromthe ADC sample rate. As also described above, the Nyquist foldedbandpass sampling receiver of FIG. 3 allows multiple Nyquist zones tofold on top of each other. Through a modulated sampling clock, separatemodulations are induced within each Nyquist zone. When the Nyquist zonesfold on top of each other, the different signals can then be separatedbased on the fact that the added modulation is different for eachNyquist zone and can be identified in the signals. Again, the analoginterpolation filter is not required for this architecture, but it cansimplify the design and provides further degrees of freedom in thereceiver design.

In some cases, it may be difficult to meet the anti-aliasingrequirements using a tunable filter for the reconfigurable direct RFbandpass sampling receiver. To overcome this potential problem, thereconfigurable direct RF bandpass sampling receiver can be combined withthe Nyquist folded receiver architecture by modulating the tunable RFsampling clock and easing the tunable anti-alias filter requirements sothat some aliasing may be allowed. The resulting receiver will havesimilar functionality as the reconfigurable direct RF receiver withsimplified anti-aliasing filter requirements. These simplifiedrequirements provide potential advantages including a reduction in cost,a reduction in insertion loss, improved noise figure, etc. While someNyquist folding will occur, the Nyquist folding will, depending upon thefront-end anti-aliasing filter utilized, be relatively minimal withperhaps only partial fold-over or perhaps only one or two additionalNyquist zones folded in. Thus the signal separation problem will bemanageable for a high-dynamic range system compared to a Nyquist foldedreceiver design with a ultra wideband front-end filter. Thisarchitecture is further explained with respect to FIGS. 9A and 9B.

FIG. 9A is a block diagram for a Nyquist folded bandpass samplingreceiver having a modulated and possibly tunable sampling clock thatcombines aspects of FIG. 1, FIG. 2A and FIG. 3. In the embodiment 900,the antenna 202 provides an RF input signal to a front-end filter 302that is implemented as an ultra wideband (UWB) filter 302 that allowsmultiple Nyquist zones to be passed through for further processing. Asdiscussed above, the signal is then applied to RF sampler 204 thatreceives a modulated RF sampling clock signal from clock circuitry 304that causes different signal modulations in different Nyquist zones. Asfurther discussed above, the modulated RF sampling clock may also behave a tunable frequency, if desired. The output of the RF sampler 204then passes to the analog interpolation filter 206. The signal is thendigitized by ADC 210, which receives a quantization clock signal fromADC clock circuitry 214. Next, the signal is processed by a DSP 902 thatis configured to unfold the Nyquist folding caused by the priorprocessing, as discussed above, and to determine the Nyquist zone fromwhich signals originated.

FIG. 9B is a block diagram for a Nyquist folded bandpass sampling with atunable and modulated sampling clock and a front-end filter thatprovides some anti-aliasing. For this embodiment 950 as compared toembodiment 900 in FIG. 9A, the front-end filter has now been implementedas a tunable anti-alias filter 912 that allows more than one Nyquistzone to pass through, but is not as wide as the front-end filter 302 inFIG. 9A. Thus, the front-end filter 912 is configured to allow morealiasing than the front-end filter 104 in FIG. 2A, but is configured toprovide more anti-aliasing than the ultra wideband filter 302 in FIG. 3.As depicted, the RF sampling clock circuitry 914 provides an RF samplingclock signal that is both modulated and tunable in frequency. Inaddition, the DSP 916 is modified from the DSP that would processsignals from the ADC 210 in FIG. 2A, so that it allows for someunfolding of the Nyquist zones, as does the DSP 902 in FIG. 9A.

As discussed above, the architecture of FIG. 9B is different from theembodiments of FIG. 1 and FIG. 2A in that the front-end filter isconfigured to allow aliasing (and limited Nyquist fold-over) in order toreduce the requirements on a front-end tunable anti-aliasing filter. Incontrast with the Nyquist folded receiver of FIG. 3, the front-endfilter 912 is a tunable anti-aliasing filter rather than a wide-openfixed ultra wideband (UWB) filter. This anti-aliasing filter 912,however, does allow some amount of aliasing. The resulting performanceof the embodiment 950 will be more similar to the reconfigurable directRF bandpass sampling receiver of FIG. 1 than the Nyquist folded receiverof FIG. 3 both in bandwidth and dynamic range. While the Nyquist foldedreceiver might have a tunable as well as modulated clock, the purpose ofthe tuning is to shift across many Nyquist zones simultaneously asopposed to avoiding or centering between Nyquist fold-over points. It isnoted that other variations and implementations could also be providedto and utilized with this architecture, if desired.

With respect to FIGS. 10A and 10B, an embodiment is discussed for usingan injected pilot tone in order to facilitate the recovery of themodulated sample clock. This approach to recover the clock signalrequires less hardware than a separate clock synchronization circuit, aswell as allowing for a more precise alignment of the RF sampling clock,because the sampling clock modulation is induced on the pilot tonewaveform and included with the sampled output data.

FIG. 10A is a block diagram for a Nyquist folded bandpass samplingreceiver 1000 utilizing an injected pilot tone to facilitate clockrecovery. The receiver architecture depicted is similar to embodiment300 of FIG. 3. A front-end filter 302 is implemented as a ultra wideband(UWB) filter that is multiple Nyquist zones wide to allow multipleNyquist zones to pass through. An RF sampler 204 then samples thesignals using a modulated RF sampling clock signal from clock circuitry304. The sampled signals are then provided to an analog interpolationfilter 206, to an intermediate frequency amplifier (IF Amp) 208, and toan ADC 210. The ADC 210 receives a quantization clock signal from ADCclock circuitry 214. In contrast to FIG. 3, the embodiment 1000 includespilot tone injection circuitry 1002 configured to inject a pilot toneinto the system such that it is combined with the RF input going to thefront-end filter 302. The pilot tone can be, for example, a signal ofknown waveform having unknown modulation phase timing. Other injectedsignals may also be used, if desired. The DSP circuitry 1004 is furtherconfigured to provide Nyquist folded processing to unfold the foldedNyquist zones. In addition, the DSP circuitry 1004 is further configuredto recover the sampling clock signal utilizing the pilot tone injectedinto the RF input.

FIG. 10B is a block diagram for example processing for DSP 1004 thatcould be used to recover the clock signal and analyze the signal data.As shown, the quantized Nyquist folded data is processed through twopaths. One path is a delay path. The other path detects the samplingclock and then provides it back to the delayed path. In particular, thequantized Nyquist folded data is provided to filter 1052 which isconfigured to be a matched digital filter that passes the RF samplingclock frequency. The output of filter 1052 is then provided to detectorand time measurement block 1054 and then to RF sample clock synthesisblock 1056 before being provided to block 1058. The quantized Nyquistfolded data is also provided to digital delay block 1060 where the datasignal is delayed by a number of clock cycles corresponding to knowndelays in the matched filter 1052, the detection processing in block1054 and the synthesis processing in block 1056. The output of thedigital delay block 1060 is provided to processing block 1058 where theRF sampling clock signal is removed from the data signal. The output ofblock 1058 is then provided to block 1062 for signal detection andmeasurement. Block 1062 also receives the synthesized clock signal fromblock 1056. In operation, block 1062 processes the data signal includingperforming modulation measurement and unfolding to the original Nyquistzone based on the known and recovered RF sampling clock. Block 1062 thenoutputs data related to the recovered signals.

In operation, a known pilot tone can be injected into a first Nyquistzone before the RF sampler 204, as shown in FIG. 10A. Because theinduced modulation in Nyquist zone X will have a ceiling (X/2) scalefactor on the RF sample clock modulation, the pilot tone injected intothe first will have an induced modulation exactly equal in amplitude tothe modulated sample clock. Because the modulated sample clock waveformis known, a matched filter can be designed to detect the modulatedsample clock. Once detected, the exact time reference of the modulatedsample clock can be measured. Because the modulated sample clock is nowembedded in the data, the measured modulated clock time reference isautomatically relative to the incoming RF data. For example, themodulated RF sampling clock can be a triangular waveform that isfrequency modulated from 2000 to 2010 Gsps with a period of 20 μsec. Theinjected pilot tone can be a continuous wave un-modulated tone at 2.155GHz. In this particular example, therefore, the pilot tone is injectedinto the second Nyquist zone. Thus, the induced frequency modulationafter Nyquist folding will equal the clock. It is noted that the knowninjected tone may be selected, as desired, for injection into the signalpath. As indicated above, the injected tone may be a continuous wavetone. It can also be an arbitrary waveform signal or any other desiredinjected tone.

FIG. 10B shows one approach to analyzing the signal data. As depicted,matched filtering is performed on the incoming RF sampled data using amatched filter 1052 based on the known RF sampling clock modulation.Then, when the RF sampling clock is detected in block 1054, theassociated time reference can be used to replicate the RF sample clockmodulation in block 1056, which is then subtracted from the input datain block 1058 and which is also used for determination of the originalNyquist zone of any remaining signals in the data in block 1062. It isalso noted that a priori information can also be applied for an improvedestimate of the originating Nyquist zone.

The recovery of the modulated clock signal provided by the exampleembodiments of FIG. 10A and FIG. 10B is advantageous. If the modulatedRF sampling clock is known exactly (e.g., waveform shape and timereference), then more flexible modulation clocks may be used withoutinducing ambiguity. For example, a linear frequency modulated (FM) sweepmay be used without knowing the time reference because the slope isconstant (except for the sweep repeats). With a sinusoidal FMmodulation, it is difficult to tell from which Nyquist zone a signalcomes without knowing the instantaneous slope of RF sampling clockfrequency modulation at that time. By recovering the modulated clocksignal, this ambiguity can be resolved.

Thus, the embodiments of FIG. 10A and FIG. 10B allow for increasedflexibility in the selection of clock modulations, and this ability toutilize a wider range of clock modulations provides advantages. In part,this flexibility allows for a wider range of signals to be processedsuccessfully and allows for increased flexibility in programming themodulation of the clock in order to make system harder to defeat bycounter measures. For example, a pulse with some linear frequencymodulation will be difficult to distinguish from a pulse in a higherNyquist zone with induced linear frequency modulation if the RF samplingclock modulation is a linear frequency modulation. However, if the RFsampling clock modulation is triangle waveform, for example, only atriangle modulation waveform that was time-aligned with the RF samplingclock would result in ambiguity (other than specially designedasymmetrical triangle modulation waveforms). If the RF sampling clockmodulation is alternated or adjusted over time during operation of thecommunication system, then no standard waveform would result inambiguity. Another benefit is that the RF sampling clock modulation doesnot have to be tightly controlled because clock can be measured ratherthan simply relying upon the expected clock signal.

Is noted that other clock synchronization circuitry could also be used.For example, another approach to acquire knowledge of the clock from thedata signal could be to include an electronic synchronization circuit toaccurately synchronize the RF sampling clock modulation with associatedtime stamping or referencing to the RF input data. This approach couldbe done for the Nyquist folded receiver, but this approach would requiresome additional hardware and would be susceptible to problems from timedelay variations. The technique in FIG. 10A and FIG. 10B does notrequire synchronization hardware, although an alternative embodimentcould be to use a synchronization circuit, resulting in a more complexhardware system, possible calibration difficulties, and simpler digitalsignal processing in the DSP. Other variations could also beimplemented, if desired.

Further modifications and alternative embodiments of this invention willbe apparent to those skilled in the art in view of this description. Itwill be recognized, therefore, that the present invention is not limitedby these example arrangements. Accordingly, this description is to beconstrued as illustrative only and is for the purpose of teaching thoseskilled in the art the manner of carrying out the invention. It is to beunderstood that the forms of the invention herein shown and describedare to be taken as the presently preferred embodiments. Various changesmay be made in the implementations and architectures. For example,equivalent elements may be substituted for those illustrated anddescribed herein, and certain features of the invention may be utilizedindependently of the use of other features, all as would be apparent toone skilled in the art after having the benefit of this description ofthe invention.

1. Receive path circuitry for a bandpass sampling receiver having foldedNyquist zones and ultra wideband pulse capabilities, comprising:wideband filter circuitry receiving an RF input signal, having a centerfrequency within a frequency range of interest and having a bandwidthless than or equal to the frequency range of interest and wide enough tocover multiple Nyquist zones associated with a modulated sampling clock;narrow band filter circuitry receiving the RF input signal and having acenter frequency within a frequency range of interest and having abandwidth less than the Nyquist bandwidth of the modulated samplingclock; and sampling circuitry configured to receive a filtered signalfrom the wideband filter circuitry, to receive a filtered signal fromthe narrow band filter circuitry, and to receive the modulated samplingclock signal as an input, the modulated sampling clock signal beingconfigured to provide non-uniform sampling for signals within themultiple Nyquist zones.
 2. The receive path circuitry of claim 1,wherein the modulated sampling clock signal comprises a frequencymodulated clock signal.
 3. The receive path circuitry of claim 2,wherein the frequency modulated clock signal comprises a linear sawtoothmodulation, a sinusoidal modulation, a triangle modulation, a frequencyshift key modulation, a frequency agile modulation, a communicationsfrequency modulation, or a combination thereof.
 4. The receive pathcircuitry of claim 2, wherein the sampling circuitry comprises:non-quantizing sampling circuitry; an analog interpolation filtercoupled to receive the output of the non-quantizing sampling circuitry,the analog interpolation filter having a center frequency within aNyquist zone of operation for the non-quantizing sampling circuitry; andanalog to digital converter (ADC) circuitry configured to receive aquantization sampling clock signal and to quantize an analog signalreceived from the analog interpolation filter.
 5. The receive pathcircuitry of claim 4, wherein the analog interpolation filter comprisesa tunable or switchable interpolation filter.
 6. The receiver pathcircuitry of claim 1, wherein the modulated sampling clock comprises afrequency modulated sampling clock having a tunable frequency or anadjustable modulation or both.
 7. The receive path circuitry of claim 1,wherein the wideband filter circuitry comprises a wideband filter havingbandwidth of 10 GHz or more, the modulated sampling clock has a samplingrate of 1 GHz or more, and the narrow band filter circuitry comprises abandwidth less than 1 GHz.
 8. The receive path circuitry of claim 1,wherein the wideband filter circuitry comprises a wideband filter havingbandwidth of 20 GHz or more, the modulated sampling clock has a samplingrate of 2 GHz or more, and the narrow band filter circuitry comprises abandwidth less than 2 GHz.
 9. The receive path circuitry of claim 1,wherein the wideband filter circuitry comprises a tunable bandpassfilter having a tunable center frequency dependent upon a filter controlsignal, the center frequency being tunable across the frequency range ofinterest, and the tunable bandpass filter having a bandwidth less thanthe frequency range of interest.
 10. The receive path circuitry of claim1, wherein the narrow band filter circuitry comprises one or moretunable narrow band filters each having a tunable center frequencydependent upon a filter control signal and each being tunable across thefrequency range of interest.
 11. The receive path circuitry of claim 1,further comprising digital signal processing circuitry coupled toreceive an output from the sampling circuitry and configured to identifyultra wideband pulses based upon signals passed through the narrow bandfilter circuitry.
 12. A method for bandpass sampling of signals usingfolded Nyquist zones, comprising: utilizing wideband filter circuitry tofilter an RF input signal within a frequency range of interest, thebandpass filter having a bandwidth less than or equal to the frequencyrange of interest and wide enough to cover multiple Nyquist zonesassociated with a modulated sampling clock; also filtering the RF inputsignal with narrow band filter circuitry having a center frequencywithin a frequency range of interest and having a bandwidth less thanthe Nyquist bandwidth of the modulated sampling clock; combining anoutput signal from the wideband filter circuitry with an output signalfrom the narrow band filter circuitry to generate a combined filteredsignal; generating a modulated sampling clock configured to providenon-uniform sampling for signals within the multiple Nyquist zones; andbandpass sampling the combined filtered signal from the wideband filtercircuitry covering multiple Nyquist zones using the modulated samplingclock.
 13. The method of claim 12, wherein the generating step comprisesgenerating a chirp frequency modulated clock signal for the modulatedsampling clock signal.
 14. The method of claim 12, wherein the bandpasssampling step comprises: bandpass sampling the combined filtered signalwithout quantizing the signal; filtering the sampled signal with ananalog interpolation filter having a center frequency within a Nyquistzone; and quantizing a signal received from the analog interpolationfilter.
 15. The method of claim 14, wherein the generating stepcomprises generating a frequency modulated clock signal for themodulated sampling clock signal.
 16. The method of claim 14, furthercomprising using a tunable or switchable interpolation filter as theanalog interpolation filter.
 17. The method of claim 12, wherein theutilizing step comprises utilizing wideband filter circuitry having abandwidth of 10 GHz or more, wherein the generating step comprisesgenerating a modulated sampling clock signal having a sampling rate of 1GHz or more, and wherein the filtering step comprises filtering withnarrow band filter circuitry having a bandwidth less than 1 GHz.
 18. Themethod of claim 12, wherein the utilizing step comprises utilizingwideband filter circuitry having a bandwidth of 20 GHz or more, whereinthe generating step comprises generating a modulated sampling clocksignal having a sampling rate of 2 GHz or more, and wherein thefiltering step comprises filtering with narrow band filter circuitryhaving a bandwidth less than 2 GHz.
 19. The, method of claim 12, whereinthe utilizing step comprises utilizing one or more tunable bandpassfilters and further comprising tuning the center frequency of thetunable bandpass filter within the frequency range of interest.
 20. Themethod of claim 12, wherein the filtering step comprises filtering theRF input signal with one or more tunable narrow band filters and furthercomprising tuning the center frequency of the one or more tunable narrowband filters within the frequency range of interest.
 21. The method ofclaim 12, further comprising processing sampled signals to identifyultra wideband pulses based upon signals passed through the narrow bandfilter circuitry.